The GaN high-electron mobility transistor (HEMT) has great potential applications in high frequency and high power electronic devices.^{[1– 3]} With the development of GaN transistor technology, the GaN HEMT has demonstrated its high power density and high efficiency over the Si- and GaAs-based radio-frequency (RF) and microwave transistors.^{[4, 5]} Thus, for the same output power, the reduction in device size can be realized using GaN-based devices in place of conventional devices. AlGaN/GaN heterostructure has the advantages of the discontinuity of conduction band Δ E and large polarization efficient in nitride, which results in a large piezoelectric polarization in the epitaxial layer of GaN.^{[6– 8]} The high concentration of two-dimensional electron gas (2DEG) is observed at the interface of AlGaN/GaN heterostructure. Therefore, AlGaN/GaN HEMT is an important tunable material for RF and microwave devices by controlling the concentration of 2DEG under gate voltage (V_g). The RF readout circuit, invented by Schoelkopf et al.^[9] in 1998, is used to investigate the RF single-electron transistor for the most sensitive and fastest electrometer.^[10] The device, such as the single-electron transistor or quantum point contact based on Si or AlGaAs/GaAs heterostructure, is embedded in an RF transmission line of the readout circuit. The changed properties of the device are reflected sensitively from the RF output signal. In China, the use of RF readout to investigate the device has been studied only a very little. In the present paper, we develop a reflective RF readout circuit in which a tunable AlGaN/GaN HEMT device is assembled in the circuit to investigate the performance of the RF readout setup. Under the applied V_g, the changes of impedance including gate capacitance and source– drain conductance are reflected sensitively to the RF output signal. The sensitive changes in resonant frequency and magnitude of output signal indicate that the RF readout setup can be applied to the high-speed detector for the applications in electromagnetic or optoelectronic devices.^[11]

The Al_0.3Ga_0.7N/GaN heterostructure provided a 2DEG about 25 nm below the surface. The electron mobility μ and density n_s at 300  K were μ = 1950  cm²/V· s and n_s = 1.4 × 10¹³  cm^{− 2}, respectively, which were determined by Hall measurement.^[12] Ultra-violet exposure, lithography, lift-off, Ohmic electrode preparation, and ion beam etching techniques were used to fabricate the HEMT microstructures. The gate length of HEMT was l = 2  μ m and the channel (mesa) width was w = 8  μ m. A thin film of 10-nm Al₂O₃ was deposited as the insulating layer by atomic layer deposition. The top electrode of the Ni/Au layer with 200  nm was fabricated by using electron beam evaporation. The microstructure of HEMT was characterized by using scanning electron microscopy (SEM).

The radio frequency (RF) readout setup mainly included a voltage-controlled oscillator (VCO), directional coupler, low noise amplifier (LNA), band-pass filter (BPF), DC block, and RF detector (Agilent 8471D). The output power from VCO (0.4  GHz– 1.3  GHz) was about + 8  dBm. HEMT and high-Q inductors (L = 100  nH) were surface mounted on the 50-Ω coplanar waveguide transmission line, which is fabricated on a high-frequency printed-circuit board (Rogers 3010). The HEMT device was electrically connected to the PCB by using short gold bond wires. The mounted inductor (L) and the capacitance (C) of HEMT formed an LC resonant unit. The LC circuit introduced a discontinuity in the electromagnetic impedance in the transmission line which reflected an incident RF signal. All RF components and transmission lines were connected by soft coaxial cables. The gate voltage V_g was supplied by a Yokogawa DC source meter.

Figure  1 shows the schematic diagram of the RF readout circuit. The incoming RF signal from the VCO is coupled to the HEMT device by a directional coupler and the reflected RF signal passes through the directional coupler to LNA, BPF, and DC block to the RF detector. The RF detector is used in RF circuits as a detecting element in the leveling loop for power monitoring. Finally, the reflected RF signal can be monitored by a multimeter. The inset shows the image of the transmission line where the HEMT device and high-Q inductors are surface mounted. The device on the copper holder must be well grounded in order to avoid damaging the HEMT device by static electricity. An SEM photo of the HEMT is also shown in the inset where mesa and Ohmic electrodes are displayed.

Fig.  1.

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	Fig.  1. Schematic diagram of the RF readout circuit and equivalent circuit of HEMT. Inset shows the image of the transmission line with the HEMT device and SEM photo of HEMT.

Figure  2 shows the variations of the transfer characteristic curve of AlGaN/GaN HEMT, i.e., drain– source DC current I_ds with gate voltage V_g. For I_ds– V_g measurement, a DC voltage V_ds is applied to the drain and V_g is applied to the gate. A multimeter through a low-noise current preamplifier with a sensitivity of 10^{− 4}  A/V is used to monitor the I_ds at source. At a fixed V_ds, I_ds decreases with the increase of | V_g| and reaches a saturation state when V_g < − 3.75  V, showing a switch characteristic of depleting the electrons under the control of V_g. The current I_ds can be described as follows:

where μ is the electron mobility, C_g is the gate capacitance, and V_th is a threshold voltage where I_ds changes abruptly. Generally, equation  (1) is used to describe the I_ds under the gate and it is difficult to fit the whole I_ds– V_g curve because the C_g is also a function of V_g. When V_g is zero, I_ds increases greatly with the increase of V_ds because the electrons in the channel of the HEMT are accelerated by V_ds, thus resulting in the enhancement of the drift current.

Fig.  2.

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	Fig.  2. Curves of Ids versus Vg at different values of applied Vds. Inset shows the schematic diagram of GaN/AlGaN HEMT.

Figure  3 shows the variations of the RF reflection signal V_RF with frequency. Two remarkable characteristics are observed. Firstly, the resonance frequency f_r shifts from 805  MHz to 816  MHz with the increase of | V_g| . Secondly, it is found that the magnitude of RF signal V_RF also decreases significantly around the resonance frequency with the increase of applied V_g. We consider that the changes of f_r and the magnitude of V_RF are strongly dependent on the impedance of HEMT under V_g, namely, the changes of gate capacitance and conductance. In an HEMT microstructure, we estimate the geometry capacitance under the gate to be C_geom = wlɛ ₀ɛ _AlGaN/d_AlGaN = 0.053  pF, where wl = 16  μ m² is the area under the gate, ɛ ₀ is the vacuum permittivity, ɛ _AlGaN = 9.4 is the relative permittivity of AlGaN, and d_AlGaN = 25  nm is the distance between the gate and 2DEG. With the increase of V_g, the electrons under the gate are gradually depleted, thus resulting in the change of electric charges Δ q in the GaN layer. The reduced electron density in 2DEG can be equivalent to the increase of distance d_AlGaN, therefore, the gate capacitance decreases with the applied V_g.^{[13, 14]} According to the resonance of the designed LC circuit, the resonance frequency is expressed as

which is mainly dominated by the capacitance C_RF including gate capacitance and parasitic capacitance, when L = 100  nH is a constant in our experiment. Therefore, f_r increases with the decrease of C_RF under V_g, which is consistent with the observed shift of f_r in Fig.  3. On the other hand, the source– drain resistance R_ds will also increase with the applied V_g increasing due to the depletion of electrons. It is suggested that the variation of the magnitude of reflection signal V_RF originates from the change of source– drain conductance.

Fig.  3.

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	Fig.  3. Variations of reflected voltage signal with frequency at different values of applied Vg.

Figure  4(a) shows the calculated capacitance C_RF as a function of V_g from Eq.  (2) by using the measured f_r where the value of the inductor is L = 100  nH. Clearly, the obtained C_RF displays the V_g-controlled characteristics, which indicates that the changed capacitance of the HEMT is reflected sensitively at V_th by the RF readout circuit. We observe that C_RF varies from 0.378  pF to 0.366 pF when the value of V_g changes from 0  V to − 4  V. The tunability τ = [C_RF(V_g = 0)– C_RF(V_g = − 4  V)]/C_RF(V_g = 0) is about 3.2% around the frequency of 816  MHz. Actually, the tunability of gate capacitance is remarkable from the measurement of the gate capacitance C_meas at a low frequency of 1  kHz by using an Agilent B1500A semiconductor device analyzer as shown in Fig.  4(b). The V_g-controlled C_meas varies from 0.25  pF to 0.042  pF, and the tunability reaches up to 83.2%, showing a good V_g-controlled performance at the low frequency. We consider that the difference in tunability mainly results from the difference in frequency response of gate capacitance.

Fig.  4.

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	Fig.  4. Variations of (a) calculated capacitance and (b) measured capacitance with Vg at 1  kHz.

Figure  5 shows the variations of source– drain conductance G(1/R_ds) and the valley of RF reflection signal V_m in Fig.  3 with V_g. The conductance of HEMT G = n_seμ w/l is mainly dependent on the electron density n_s and mobility μ when the micro-structure of HEMT is fixed. With the increase of applied V_g, conductance G decreases abruptly in the vicinity of V_th due to the depletion of electrons. We find that the V_m also exhibits the same variation trend as the G– V_g relationship. From the RF equivalent circuit in our previous article, ^[10] the reflection coefficient is Γ = (Z_in − Z₀)/(Z_in + Z₀), where Z_in is the input impedance and Z₀ = 50  Ω is a characteristic impedance. The input impedance can be expressed as Z_in = j2π fL + R_ds/[1 + j2π fR_dsC] + Z₀. In the experiment, capacitance C mainly includes gate capacitance C_g and the parasitic capacitance of the circuit. The sensitivity of the output signal under the gate voltage is proportional to both d(1 − Γ )/dR_ds(V_g) and dR_ds(V_g)/dV_g. The first term is determined by the characteristics of the RF transmission line, while the second term is determined by the characteristic of the HEMT. The intrinsic sensitivity | dR_ds/dV_g| of the readout circuit is strongly dependent on the rate of changed drain– source resistance R_ds(V_g). Therefore, the design and fabrication of good performance of HEMT with large | dR_ds/dV_g| are the key to improving the sensitivity of the readout setup, namely, the rapid change of source– drain resistance near threshold voltage V_th under V_g. In the present experiment, the sensitivity of the readout setup can increase up to 5.6× 10⁸. When a threshold voltage V_th is applied to the gate of the HEMT, the RF readout setup can be used to detect sensitively the change of impedance of HEMT under the external radiation, such as terahertz radiation or light.

Fig.  5.

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	Fig.  5. Variations of the valley of RF reflection signal Vm and conductance with Vg.

We designed a reflective RF readout circuit and investigated the performance of the HEMT-based readout setup. The RF setup is used to detect the changed impedance of the HEMT by analyzing the resonance frequency and the magnitude of the RF reflection signal. It is suggested that the designed RF readout setup can be applied to the readout circuit for high-sensitivity optic– electric detection.